1. Field of the Invention
The invention relates t(c) a method for controlling a three-phase converter with a voltage intermediate circuit by use of pulse-width modulation for supplying a polyphase system, in particular a three-phase alternating-current system. The polyphase system may likewise be a machine which is operated as a motor or as a generator (three-phase machine), such as a three-phase power supply system. In particular, the invention also relates to a three-phase motor of a domestic washing machine or of a drive for art automobile. The three-phase motor can be operated in the field-weakening mode or constant-power mode at full or almost full motor voltage. In particular, the converter which is provided for supplying a three-phase power supply system can be fed by a solar generator. In this respect, the invention is also suitable for operating a photovoltaic system or for operating a system having a different DC power source.
A three-phase converter for generating the output voltages for a polyphase system from a voltage intermediate circuit according to FIG. 1 has long been known from the prior art. A converter of this kind contains three half-bridges with in each ease two pairs containing a switching element and a diode which is connected in parallel. Contact is made with the individual phases for the polyphase system between the pairs of a half-bridge in each case. The voltages required for the polyphase system are given by the potential differences between the individual phases in relation to one another. The respective phase is alternately clamped to the upper or the lower intermediate circuit potential by alternately switching the two switching elements of a half-bridge.
Driving the, in total, six switching elements, of which in each case two are distributed to one of the three half-bridges of the converter, by pulse-width modulation is also known, reference being made to Handley, P. G.; Boys, J. T.; “Practical Real-Time PWM Modulators: An Assessment”, IEE Proceedings B, Electric Power Applications, Volume 139, pages 96 et. seq., Issue 2, March 1992. Modern converters almost exclusively exhibit switchable power semiconductors as switching elements. In addition to field-effect transistors, MOSFETs (“Metal Oxide Semiconductor” —field-effect transistors) and bipolar transistors, in particular so-called IGBTs (Insulated Gate Bipolar Transistors), are used, as power semiconductors of this kind. In this case, the latter combine the advantages of field-effect and Bipolar transistors.
In the event of pulse-width modulation, the potential or the voltage in relation to a reference potential, for example Ua in accordance with FIG. 1, is set for the phase of a half-bridge by changing the switching states of the corresponding switching elements within a predefined period duration. In other words, the duty ratio is modulated at a constant frequency in the event of pulse-width modulation. The value of the output voltage of a period duration is the result of “summation” of the switching states set in each case or results from the ratio of the switching times of the clamping to the upper intermediate circuit potential and the clamping to the lower intermediate circuit potential.
Permissible switching states of the converter are, for in each ease one half-bridge, “TopOn”, that is to say upper switching element on and lower switching element off, “Bot(-tom)On”, that is to say tower switching element on and upper switching element off, and “dead time” if both of the switching elements are switched off. The last switching state of a “dead time” is usually set only for negligibly small time periods (approximately 1% duration) between the states “TopOn” or “BotOn”. Therefore, in the event of a negligible dead time within a period duration Tp (for example 100 ms), “TopOn” for the time Ttop and then “BotOn” for the time Tbot=Tp−Ttop are alternately set. If the voltage of the respective half-bridges Ua, Ub and Uc according to FIG. 1 tends toward the tower branch of the voltage intermediate circuit, the voltage Udc being applied between the branches of the voltage intermediate circuit, the result is a pulse control factor of the respective half-bridges of T=Ttop/Tp. The mean voltage for the half-bridge a is then given, by way of example, byUa=τa·Udc.
During stationary operation of the polyphase system, the setpoint potential differences which are to be generated or the setpoint voltages between the three phases are generally sinusoidal in three phases. In the case of a three-phase alternating current, the phase difference between the individual phases is 120° in each case. The pulse control factors of the three half-bridges of the converter are then generated by a rule of the following kind:
            τ      a        =                            τ          ^                ·                  cos          ⁡                      (                          ω              ⁢                                                          ⁢              t                        )                              +              τ        0                        τ      b        =                            τ          ^                ·                  cos          ⁡                      (                                          ω                ⁢                                                                  ⁢                t                            -                                                2                  ⁢                  π                                3                                      )                              +              τ        0                        τ      c        =                            τ          ^                ·                  cos          ⁡                      (                                          ω                ⁢                                                                  ⁢                t                            -                                                4                  ⁢                  π                                3                                      )                              +              τ        0            where {circumflex over (τ)}=ûref/Udc is the amplitude of the drive level, ûref is the amplitude of the setpoint fundamental, Udc is the voltage of the DC intermediate circuit and τ0 is the common offset of the pulse control factors.
Identifiably different variants are available for generating the desired voltages between the individual phases of the converter Uab=Ua−Ub, Ubc=Ub−Uc, Uca=Uc−Ua by pulse-width modulation. Since only the voltages or difference signals between the phases Uab, Ubc, Uca are required in sinusoidal form in a polyphase system, a common offset U0, also called a zero system, can be superimposed on the individual potentials Ua, Ub, Uc. The zero system U0 is selected such that the mean individual potentials at the converter branches are in the range of between zero and Udc, that is to say τ0 is selected such that the pulse control factors τa, τb and τc are between zero and one. In this case, a pulse control factor of τa=0 means that the corresponding converter branch a is not switched but rather is permanently clamped to the lower intermediate circuit potential during the entire period duration. A pulse control factor of τa=1 means that the corresponding converter branch is not switched but rather is permanently clamped to the upper intermediate circuit potential during the entire period duration.
Consequently, it can be seen that, in particular for a three-phase sinusoidal system, it is possible, in the difference signal between the phases, to generate the individual potentials of the phases by pulse-width modulation in such a way that in each case only two half-bridges are switched, while in each case the third half-bridge is permanently clamped to the upper or to the lower intermediate circuit potential and the switch-on period of this half-bridge is either one or zero. The modulation method of two-phase docking exhibits lower switching losses in comparison to the modulation method in which all three half-bridges are operated in a clocked manner in order to generate the individual potentials. The switching frequency is reduced.
A modulation method of two-phase switching is known, for example from published; European patent application EP 036 514 A1, corresponding to U.S. Pat. No. 4,333,948. In the document, only two phases or two of the half-bridges are switched in respect of each of the periods of the desired inusoidal output signal for in each case 60°, while in each case the third phase or half-bridge is permanently clamped to the upper or the lower intermediate circuit potential. As a result, the switching losses in the switching elements are reduced in comparison to the modulation method of three-phase switching. The switching frequency is reduced by a third. However, one disadvantage is that the line losses in the permanently clamped switching elements increase.
In another method of pulse-width modulation for a three-phase converter, so-called space-vector modulation, the individual possible converter switching states are assigned vectors which specifically indicate the switching state given in each case by way of a set of three numbers. The above-described switching states “TopOn” and “BotOn” of a respective half-bridge are assigned the numerical values “1” or “0” in the process. The half-bridges a, b and c correspond to the first, second and, respectively, third positions in the set of three numbers. Therefore, tor example, the set of three numbers (1,1,0) describes the switching state of the converter, with the first half-bridge a and the second half-bridge b exhibiting the switching state “TopOn” and the third half-bridge c exhibiting the switching state “BotOn”. The sets of three numbers (0,0,0) and (1,1,1) are also called zero vectors since all the phases of the converter are either clamped to the upper or to the lower intermediate circuit potential in these switching states. The voltages or potential differences between the individual phases are therefore zero, and therefore these vectors have no relevance to the polyphase system. In this respect, the zero vectors are inactive states of the converter. Therefore, a total of six active possible switching configurations remain available in the case of a three-phase converter.
In respect of the space-vector modulation method, the vectors which are assigned to the active switching states are applied in accordance with FIG. 2, with adjacent vectors in each case differing only by one switching state of a half-bridge. The sectors which span the space between two adjacent vectors in each case are accordingly continuously numbered. In mathematical terms, the space-vector modulation method according to FIG. 2 is a transformation of the three-dimensional description of the output variables of a three-phase converter into the two-dimensional space. The output voltages of the converter rotate as two-dimensional vectors within the hexagon according to FIG. 2 which is spanned by the six base vectors of the states of the converter.
Voltage vectors between in each case two base vectors or active states of the converter are generated by the switch-on periods of the active states. Switching states of the zero vectors are added within the period duration, in order to generate voltage vectors which do not reach as far as the edge of the hexagon. The switch-on times of all the selected switching states add up to form the period duration of the pulse-width modulation. If no zero vectors are added, only output voltages whose associated vectors in the diagram according to FIG. 2 end at the edge of the hexagon can be generated.
Generating a voltage vector with intelligent connection of the zero vectors corresponds to the above-described modulation method of two-phase clocking. In each sector, the voltage values are set by adjacent base vectors or switching states of the converter which in each case differ only by one switching state of a half-bridge. For example, the base-vectors (1, 1, 0) and (1, 0, 0) which span the sector 1 differ only by the switching state of the second half-bridge. If a changeover is consequently made to a zero vector in each case, that is to say either to the zero vector (1, 1, 1) or to the zero vector (0, 0, 0) in the present case, within a sector, either the first or the third haft-bridge additionally remains permanently clamped to the upper or to the lower intermediate circuit potential.
FIG. 2 also shows that voltage vectors which end outside the inscribed hexagon cannot be generated. The maximum voltage magnitude prespecified by the base vectors cannot be reached within the sectors. Therefore, in accordance with FIG. 2, the vectors of the voltage limit which can be set with the converter end on a circle which inscribes the hexagon. If voltage values which lie outside the hexagon are required at the output terminals during driving of the converter, the voltage values are limited to the so-called hexagon limit, that is to say the corresponding vectors end on the edge of the hexagon. This is referred to as so-called overmodulation. The actual required voltage or potential profile of a phase can no longer be generated. The voltage between two phases of the converter then has a deformed sinusoidal profile, this being associated with undesired harmonics in the signal.
In the case of open-loop or closed-loop control of a converter for a polyphase system by space-vector-modulation, the variables which describe the system, and in particular the output variables of the converter, are generally transformed into a two-dimensional coordinate system. In the event of so-called field-oriented control, the two-dimensional coordinate system for describing the space-vector variables rotates with the magnetic flux of the polyphase system. The coordinates of the space-vector variables in the case of field-oriented control are denoted by d and q. Another possibility is transformation into a two-dimensional fixed-stator coordinate system. The coordinates of the space-vector variables in this system are denoted by α and β. For the purpose of controlling a polyphase system by space-vector modulation, ready measured state variables such as terminal voltages or motor currents are measured and mathematically transformed into the respective space-vector-variables, and the converter is driven to the desired switching states for generating the voltages between the phases in accordance with the transformed space-vector variables. In this case, reference is made, in particular, to the two-dimensional illustration for generating the output voltages according to FIG. 2.
In the case of current-controlled regulation of the polyphase system, the output voltages of the converter are driven in dependence on a deviation of the nominal current from a setpoint current, for example by prespecifying a setpoint value for the field-forming current component id and a setpoint value for the torque-forming current component iq. In this case, regulation of a polyphase system within the d, q coordinate system has the advantage the current component id which leads to a reactive power, because it is field-forming, and the torque-forming component iq are immediately apparent.
Driving a converter by pulse-width modulation using space-vector variables is described, for example, in U.S. Pat. No. 6,819,078 B2. To this end, a computation algorithm is specified, with which the pulse control factors τa, τb, τc can be calculated with simple computation operations from a setpoint voltage vector in the case of vector-controlled converters. Modulation methods both with three-phase switching and also with two-phase switching are possible with the specified algorithm. In this case, overmodulation is identified by a negative switch-on period for a zero vector occurring within the algorithm, this corresponding to the requirement of a voltage value outside the hexagonal limit. In the case of overmodulation, the space-vector variable which is associated with the voltage value is rescaled, and as a result the voltage values again lie within the hexagonal limit. Therefore, overmodulation with the known disadvantages is prevented.
Furthermore, European patent EP 0 840 441 B1 discloses field-oriented control of a three-phase machine, with the output voltages being controlled at the voltage ceiling. In order to provide a voltage limit, in the event of control in the d, q coordinate system, the magnitude of the torque-forming current component iq is reduced when the voltage component Ud in the flux direction reaches a limit value, and the field-forming current component id is reduced when the voltage component Uq perpendicular to the flux direction reaches a further limit value. The corresponding phase voltages of the converter are controlled in accordance with the limited voltage components Ud, Uq. In particular, the maximum available output voltage of the converter is selected for the limit values, and therefore the pulse control factors of the half-bridges, which pulse control factors are prespecified for the converter, are less than “1” and the converter is driven within the hexagonal limit. This creates a voltage reserve which is required for loading, the field-forming and/or the torque-forming current component being reduced for this purpose. Since the converter is operated below the hexagonal limit, overmodulation is avoided, this possibly leading to control problems due to voltage values which cannot be realized being required.
Published, non-prosecuted European application EP 2 192 682 A1 proposes setting the voltage requirement of a three-phase machine by a characteristic map for the field-forming current id, so that the quadrature current controller and the in-phase current controller do not reach a setting limit. If the actuating variables of the current controllers were to be limited by the maximum possible voltage, the setpoint current components id, iq would no longer be able to be controlled and the behavior of the machine would be uncertain. Overmodulation is therefore deliberately avoided here too.